Control device of synchronous electric motor, integrated motor system, pump system, and positioning system

ABSTRACT

A control device of a synchronous electric motor includes: the synchronous electric motor with three-phase stator windings Y-connected; a detection unit that detects a neutral point potential which is a potential at a Y connection point; and an inverter that drives the synchronous electric motor. The control device of the synchronous electric motor which controls the synchronous electric motor using the inverter, includes a measurement mode in which the neutral point potential is detected in a state in which the synchronous electric motor is energized by an AC current and controls the synchronous electric motor based on a value of the neutral point potential detected in the measurement mode.

TECHNICAL FIELD

The present invention relates to an electric motor driving technologyapplied to applications requiring, for example, rotational speed controlof a pump, a fan, a compressor, and a spindle motor, positioning controlof a conveyor, an elevator, and a mechanical device, and torque controlsuch as electric assist, and the like.

BACKGROUND ART

For example, in fields such as home appliances, industries, andautomobiles, a motor driving device for rotational speed control of afan, a pump, and a compressor, torque control of electric powersteering, and positioning control of a transport machine, an elevator,and the like are used. In the motor driving device, a permanent magnettype synchronous electric motor (hereinafter, referred to as “PMmotor”), which is a small-sized and highly efficient alternating current(AC) electric motor, is widely used. However, information on a magneticpole position of a rotor of the PM motor for controlling driving of thePM motor is required, so that a position sensor such as a resolver or ahall IC is indispensable. Recently, without using the position sensor,sensorless control for controlling rotation speed and torque control ofthe PM motor is spread.

By realizing the sensorless control, it is possible to reduce a cost ofthe position sensor (cost of the sensor itself, cost of wiring of thesensor, and cost of attaching and adjusting the sensor) and as thesensor becomes unnecessary, the device can be miniaturized and can beused in a bad environment.

Currently, the sensorless control of the PM motor adopts a method ofdirectly detecting the induced voltage (speed electromotive voltage)generated by rotation of the rotor and driving the PM motor by positioninformation of the rotor or a position estimation technology ofestimation-operating the rotor position from a mathematical model of thePM motor.

The method is based on a method using the speed electromotive voltage inprinciple, so that it is difficult to be applied to a region where thespeed electromotive voltage becomes small, such as a stop range, a lowspeed region, or the like. Therefore, these technologies are mainlyapplied to the speed range above a medium to high speed range and openloop control such as V/F constant control is used in the low speedregion. In a case of the open loop control, since the torque generatedby the motor cannot be freely controlled, controllability in the lowspeed region is poor and efficiency also deteriorates.

Regarding to the problem, a method of obtaining the rotor positioninformation from the low speed region is proposed.

In PTL 1, a pulse voltage is applied to two phases of the PM motor amongthree phases to detect the open voltage of the remaining one phase notenergized, so that position information is obtained from the voltage.Since the electromotive voltage of the open phase is generated accordingto the rotor position of the PM motor, the electromotive voltage can beused for estimation of the rotor position. The electromotive voltage isa voltage generated by slight inductance change in the motor by arelationship between a permanent magnet-magnetic flux attached to therotor of the PM motor and a current energized by the pulse voltage, sothat the electromotive voltage can be observed in a stop state. Thisvoltage is called “magnetic saturation electromotive voltage”.

In addition, in this method, since the electromotive voltage of thenot-energized phase (open phase) is observed, among the three phases, a120-degree energization driving for selecting and energizing the twophases is indispensable. For position-sensorless driving, it is requiredthat these energized phases are switched according to the position ofthe rotor. For switching these energized phases, “magnetic saturationelectromotive voltage” generated in the open phase is used.

The magnetic saturation electromotive voltage is changed to bemonotonously increased or decreased according to the position of therotor. In PTL 1, “threshold value” is provided in the open phaseelectromotive voltage. When the magnetic saturation electromotivevoltage reaches the threshold value, position-sensorless control isperformed by switching to the next energization phase. At that time,“threshold value” is a very important setting factor. The thresholdvalue has subtle variations for each of phases or phase windings of themotor and it is necessary to set threshold value appropriately. A methodof automatically executing adjustment work for each of motors isdescribed in PTL 2.

In PTL 2, with respect to the method described in PTL 1, by executing anautomatic threshold value adjustment routine in advance, it becomesunnecessary to manually adjust the threshold value by an operator andstart-up work of a system is saved.

Although the 120-degree energization driving is premised in theserelated arts, a sinusoidal wave driving method is already reported. InPTLs 3 and 4, by observing a connection point potential (referred to asneutral point potential) of a Y-connected three-phase windings using thethree-phase stator windings Y-connected as the PM motor, the rotorposition is estimated.

Since it is unnecessary to observe the open phase in the same manner asin PTL 1, it is possible to energize three phases at the same time andto drive the PM motor with an ideal sinusoidal current. However,detection of the neutral point potential is indispensable.

In PTL 3, a method of inserting a voltage pulse for observing theneutral point potential is described. In addition, in PTL 4, it ispossible to immediately estimation-operate the rotor position by using avoltage applied to the inverter for driving the PM motor and observingthe neutral point potential by interlocking with a PWM pulse when apulse width is modulated.

CITATION LIST Patent Literature

PTL 1: JP-A-2009-189176

PTL 2: JP-A-2012-10477

PTL 3: JP-A-2010-74898

PTL 4: Pamphlet of International Publication No. WO 13/153657

SUMMARY OF INVENTION Technical Problem

In the technology of PTL 1, it is possible to generate torque withoutstep out in a state in which the motor is stopped or in a low speed. Inaddition, PTL 2 describes automatic adjustment of “threshold value”which is an important setting number for realizing sensorless driving inPTL 1. However, since the 120-degree energization driving is the basisin both of the methods of PTLs 1 and 2, a current harmonic wave when PMmotor is driven is extremely large. As a result, there is a case where aharmonic loss may be increased and vibration/noise due to torquepulsation may be problematic. Driving with the sinusoidal current isdesirable for driving the PM motor.

In PTLs 3 and 4, it is possible that the neutral point potential of thestator winding of the PM motor is observed to drive the PM motor from azero speed by the sinusoidal current. In addition, there is nostructural restriction (for example, limitation such as being limited toembedded magnet type) of the PM motor and versatility is also high.However, these PTLs 3 and 4 have the following problems which are notsolved.

Although PTL 3 describes a method of switching energization phases ofthe three phases by using the observed neutral point potential, PTL 3does not describe a case where the neutral point potential for switchingis specifically set, a case where specifications of motors are differentfrom one another, or a case of unbalance of the three phases. Further,there is a case where the neutral point potential changes due to atorque current by magnetic circuit characteristics of the motor, but PTL3 does not describe countermeasures for the change. For this reason, inorder to realize the method of PTL 3, there is a problem in practicaluse such as an adjustment work for each of the motors and an increase ina position estimation error with respect to a load torque.

In PTL 4, the neutral point potential in each of voltage patterns isobserved when two types of voltage patterns are applied and the rotorposition of the PM motor can be estimation-operated by this signalprocess. However, in a case where it is impossible to deal withthree-phase unbalance, for example, only inductance of a specific phaseis different from the others, a large pulsation component is generatedat the estimated rotor position. In addition, although the two types ofthe voltage patterns can be created by pulse width modulation usingordinary triangular wave carrier, it is necessary to abundantly preparean AD converter, a timer, or the like as a function of a controller soas to detect the neutral point potential corresponding to each of thevoltage patterns. In a case of using an inexpensive microcomputer, thesefunctions are insufficient and the technology of PTL 4 cannot be appliedas it is. In addition, in the same manner as in PTL 3, in a case wherethe neutral point potential is changed due to the torque current, thereis a possibility that a position estimation error increases and torqueperformance deteriorates.

Solution to Problem

According to the present embodiment, a control device of a synchronouselectric motor includes: the synchronous electric motor with three-phasestator windings Y-connected; a detection unit that detects a neutralpoint potential which is a potential at a Y connection point; and aninverter that drives the synchronous electric motor. The control deviceof the synchronous electric motor which controls the synchronouselectric motor using the inverter includes a measurement mode in whichthe neutral point potential is detected in a state in which thesynchronous electric motor is energized by an AC current. It ispreferable that the control device of the synchronous electric motorcontrols the synchronous electric motor based on a value of the neutralpoint potential detected in the measurement mode.

Advantageous Effects of Invention

Since the invention is configured as described above, the followingeffects are obtained. According to the invention, since a relationshipbetween the neutral point potential of the PM motor and the rotorposition and a relationship between the torque current and the neutralpoint potential can be obtained in advance, sensorless driving of amotor with any magnetic circuit characteristics in a low speed regioncan be realized by a simple adjustment algorithm.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a diagram illustrating a configuration of a control device ofa synchronous electric motor.

FIG. 2(a) is a diagram illustrating a vector display of an outputvoltage of an inverter on αβ coordinates.

FIG. 2(b) is a diagram illustrating a relationship with rotor positionsθd of a PM motor.

FIG. 3(a) is a diagram illustrating a generation principle of a neutralpoint potential of the PM motor according to a first embodiment.

FIG. 3(b) is a diagram illustrating a generation principle of a neutralpoint potential of the PM motor according to the first embodiment.

FIG. 4 is a waveform diagram illustrating a measurement example of theneutral point potential according to the first embodiment.

FIG. 5 is a waveform diagram illustrating linearization of the neutralpoint potential according to the first embodiment.

FIG. 6 is a waveform diagram illustrating all types of changes of theneutral point potential according to the first embodiment.

FIG. 7 is a waveform diagram illustrating a change of the neutral pointpotential used for sensorless driving according to the first embodiment.

FIG. 8 is a waveform diagram illustrating a change by current dependenceof the neutral point potential used for the sensorless driving accordingto the first embodiment.

FIG. 9(a) is a diagram illustrating a relationship between a magneticflux and a torque current of an inside of the PM motor during normaldriving.

FIG. 9(b) is a diagram illustrating a relationship between a magneticflux and a torque current of the inside of the PM motor according to thefirst embodiment.

FIG. 10 is a diagram illustrating a waveform when current dependence isobtained in a measurement mode according to the first embodiment.

FIG. 11 is a flowchart illustrating an algorithm during the measurementmode according to the first embodiment.

FIG. 12(a) is a diagram illustrating a position of a voltage command V*which can output a voltage vector VA.

FIG. 12(b) is a diagram illustrating a voltage pulse and a neutral pointpotential of each of phases during a non-pulse shift.

FIG. 12(c) is a diagram illustrating a voltage pulse and a neutral pointpotential of each of phases during a pulse shift.

FIG. 13 is a block configuration diagram of a position estimatoraccording to the first embodiment.

FIG. 14 is a waveform diagram illustrating an appearance of a variationof a neutral point potential used for sensorless driving according to asecond embodiment.

FIG. 15 is a diagram illustrating switching of a coefficient tableaccording to the second embodiment.

FIG. 16 is a waveform diagram illustrating linearization of a neutralpoint potential according to a third embodiment.

FIG. 17 is a flowchart illustrating an algorithm during a measurementmode according to the third embodiment.

FIG. 18 is a flowchart illustrating an algorithm during a measurementmode according to a fourth embodiment.

FIG. 19 is a diagram illustrating a configuration of an integral motordriving system according to a fifth embodiment.

FIG. 20 is a diagram illustrating a configuration of a hydraulic pumpsystem according to a sixth embodiment.

FIG. 21 is a diagram illustrating a configuration of the hydraulic pumpsystem in which a relief valve is removed according to the sixthembodiment.

FIG. 22 is a block diagram illustrating a configuration of a positioningcontrol system according to a seventh embodiment.

DESCRIPTION OF EMBODIMENTS

Hereinafter, embodiments of the invention will be described withreference to drawings. In the embodiment of the invention, in view of aproblem in that a neutral point potential fluctuates also due to atorque current, there is provided a control device of a synchronouselectric motor which automatically adjusts a magnetic saturationcharacteristic of each of motors to be controlled, dependence of thetorque current, a three-phase imbalance characteristic, and the like andrealizes high torque sinusoidal wave driving without using a rotorposition sensor in a vicinity of zero speed. As a result, it is possibleto drive high torque with less vibrations and noises than the previouslydisclosed method. In addition, in actual driving after adjustment,position estimation-operation by a simple algorithm becomes possible, sothat the position estimation-operation can be realized by an inexpensivemicrocomputer.

First Embodiment

A control device of an alternating current (AC) electric motor accordingto the first embodiment of the invention will be described using FIGS. 1to 13.

The device is for driving a three-phase permanent magnet synchronouselectric motor 4 (hereinafter, referred to as PM motor 4) and is broadlyconfigured to include an Iq* generator 1, a controller 2, an inverter 3,and the PM motor 4 which is a driving target. The inverter 3 includes adirect current (DC) power supply 31, an inverter main circuit 32, agatedriver 33, a virtual neutral point potential generator 34, and a currentdetector 35.

Although the present embodiment uses the PM motor as an example of thedriving target, as long as an electric motor has a magnetic saturationcharacteristic with respect to a rotor position, other types of ACelectric motors can be also applied to the driving target.

The Iq* generator 1 is a control block which generates a torque currentcommand Iq* of the PM motor 4 and corresponds to an upper controller ofthe controller 2. For example, the Iq* generator 1 functions as a speedcontroller which controls a rotation speed of the PM motor 4 or a blockwhich operates a torque current command required from a state of a loaddevice such as a pump and gives the command to the controller 2.

The controller 2 is a controller which controls a vector of the PM motor4 without a rotor position sensor. The controller 2 has functions asboth of “actual operation mode” for realizing a normalposition-sensorless control and “measurement mode” for automaticallyperforming adjustment operation to each of the PM motors before theactual operation. The controller 2 switches between operation of theactual operation mode and operation of measurement mode by a switchinside the block.

The controller 2 includes switches SW21 a to SW21 e which switch betweenthe actual operation mode and the measurement mode. In the actualoperation mode, the switches SW21 a to SW21 e are switched to “1” and inthe measurement mode, the switches SW21 a to SW21 e are switched to “0”.In the actual operation mode, position estimation based on the neutralpoint potential and a vector control system using current control of adq axis are realized. In the measurement mode, a parameter required by aposition estimator 15 during the actual operation mode is obtained by analgorithm described below.

The controller 2 includes a current reproducer 11 which reproducesthree-phase alternating currents Iuc, Ivc, and Iwc based on a direct buscurrent of the inverter 3. In the present embodiment, the phase currentis reproduced by detecting the direct bus current by the currentdetector 35, but a phase current sensor may be directly used. A detailedexplanation for operation of the current reproducer 11 will be omitted.

The reproduced three-phase alternating currents Iuc, Ivc, and Iwc areconverted to a value (Id, Id) of a dq coordinate axis which is a rotorcoordinate axis of the PM motor 4 by a dq converter 12. A d axis currentId is input to a d axis current controller IdACR 7 via an adder 6. A qaxis current Iq is input to a q axis current controller IqACR 8 via theadder 6.

The d axis current controller IdACR 7 performs the current control basedon the d axis current Id and an excitation current command Id* to the PMmotor 4. The Id* which is input when the current control is performed isswitched by the switch SW21 c. In the actual operation mode, a signalfrom an Id* generator 5 a is used. In the measurement mode, a signalfrom an Id* generator 5 b is used.

The q axis current controller IqACR 8 performs the current control basedon the q axis current Iq and the q axis current command Iq*. The Iq*which is input when the current control is performed is switched by theswitch SW21 d. In the actual operation mode, a signal from an Iq*generator 1 is used. In the measurement mode, a zero generator 19 setsthe Iq* to zero.

During the actual operation mode, a d axis voltage command Vd* outputfrom the d axis current controller IdACR 7 and a q axis voltage commandVq* output from the q axis current controller IqACR 8 are input to a dqinverse converter 9. The dq inverse converter 9 converts the voltagecommands Vd* and Vq* on the dq axis into three-phase AC voltage commandsVu0, Vv0, and Vw0. Then, a pulse width modulation (PWM) unit 10generates a gate pulse signal which drives the inverter 3 based on thethree-phase AC voltage command.

Determination of conversion phase used by the dq inverse converter 9 andthe dq converter 12 will be described.

The controller 2 includes a neutral point potential amplifier 13. Theneutral point potential amplifier 13 amplifies and detects a neutralpoint potential Vn of the PM motor 4 based on a virtual neutral pointpotential Vnc of the virtual neutral point potential generator 34. Inthe present embodiment, the neutral point potential is detected based onthe virtual neutral point potential Vnc of the virtual neutral pointpotential generator 34, but a potential as a standard can bepredetermined. For example, the neutral point potential also can bedetected using another reference potential as a reference such as aground level of the DC power supply 31 of the inverter 3 or the like.

A sample/holder 14 samples/holds the detected neutral point potentialand captures the neutral point potential inside the controller. Then,the position estimator 15 estimation-operates the rotor position θdc ofthe PM motor 4 based on the detected neutral point potential Vn0(actually one of VnA to VnF). During the actual operation mode, theestimated rotor position θdc is input to the dq inverse converter 9 andthe dq converter 12. In addition, a speed operator 16 estimates a rotorspeed ω1 based on the estimated rotor position θdc.

As described below, the control device of the present embodiment hascharacteristics of energizing an AC current to the PM motor 4 andobtaining current dependence of the neutral point potential Vn at thattime. As a configuration for the characteristics, the control device ofthe AC electric motor according to the present embodiment includes an ACcommand generator 20 which applies an AC voltage Vh* to the dq inverseconverter as a q axis voltage during the measurement mode.

The neutral point potential Vn0 detected during the measurement mode(that is, energization of the AC current) is input to an estimationparameter setter 18 by the switch SW21 a. An estimation parameter setter18 sets a parameter required for position estimation during the actualoperation mode based on the neutral point potential Vn0.

In addition, the controller 2 includes a phase setter 17 which forciblymoves the rotor position to a predetermined position during themeasurement mode During the measurement mode, the conversion phase inputto the dq inverse converter 9 and the dq converter 12 is switched to asignal ouput from the phase setter 17 by the switch SW21 b.

Then, the control device of the AC electric motor according to thepresent embodiment includes the AC command generator 20 which appliesthe AC voltage Vh* to the dq inverse converter as the q axis voltageduring the measurement mode. Accordingly, current dependence of theneutral point potential is obtained by generating the AC current.

Next, a principle of position-sensorless driving based on the neutralpoint potential will be described with reference to FIGS. 2 to 7.

FIG. 2(a) is a diagram illustrating a vector display of an outputvoltage of the inverter on αβ coordinates. In FIG. 2(a), a numericalvalue “1, 0, 0” of V(1, 0, 0) indicates a switching state of U, V, and Wphases of the inverter main circuit, “1” means turning on of an upperelement, and “0” means turning on of a lower element For example, V(1,0, 0) means the states in which the upper element of the U phase isturned on and the lower elements of the V phase and W phase are turnedon. The output voltage of the inverter 3 becomes voltage patterns ofeight types which are a sum of two zero vectors (V(0, 0, 0) and V(1, 1,1)) and six non-zero vectors (VA to VF) according to the switching stateof each of three-phase switching devices (Sup to Swn).

The inverter generates a sinusoidal pulse pattern using these eightvoltage vectors. For example, for an arbitrary voltage command V*,assuming that the command is in a region 3 in FIG. 2(a), a voltagecorresponding to the V* is generated by combining vectors VB and VC anda zero vector surrounding the region 3.

FIG. 2(b) is a diagram illustrating the rotor position θd of the PMmotor on the αβ coordinates. The θd is normally defined in acounterclockwise direction based on an a axis (equal to a U-phase statorwinding position).

FIG. 3(a) illustrates a stop point potential VnA when the voltage vectorVA is applied and FIG. 3(b) illustrates a stop point potential VnD whenthe voltage vector VD is applied. The neutral point potential isrepresented by equations illustrated in FIGS. 3(a) and (b) based on avirtual neutral point potential. In this way, each of the neutral pointpotentials is observed as a divided potential of stator windings Lu, Lv,and Lw. If an inductance of each of the windings is equal with oneanother, the neutral point potential is completely zero. However, sincea magnetic flux of the rotor actually affects the winding, theinductance has a change according to the rotor position. In thefollowing description, VnA, VnB, VnC, VnD, VnE, and VnF are respectivelyused as names of the neutral point potentials generated when the voltagevectors VA, VB, VC, VD, VE, and VF are applied.

FIG. 4 illustrates a result of observing the neutral point potential byapplying a voltage vector to the actual PM motor. Both of V_(nA) andV_(nD) have changes according to the rotor position. By using dependenceof the neutral point potential for the rotor position, the rotorposition can be estimated. In the present embodiment, the rotor positionis estimation-operated using a part of the waveform of FIG. 4. FIGS.5(a) to 5(c) are illustrated as examples of linearization focusing onthe change of VnA in FIG. 4.

FIG. 5 (a) illustrates a case where a change of VnA from −60 [deg] to 0[deg] is regarded as a straight line and is made into a function asVn=Fa (θd) . When the position is estimated, it is possible to estimatethe rotor position θd from Vn by using θd=Fa⁻−(Vn).

In order to realize this, two neutral point potentials with respect tothe θd may be obtained. For example, the rotor is moved to −60 [deg] andthe VA is applied to a position of the rotor to obtain the neutral pointpotential VnA0. Then, the rotor is moved to 0 [deg] and the VA isapplied to a position of the rotor to obtain the neutral point potentialVnA1.

FIG. 5(b) is another example of linearization different from FIG. 5(a).As illustrated in FIG. 5(b), it is also possible to approximate at apredetermined point in a range from −60 [deg] to 0 [deg].

Basically, the neutral point potential Vn0 (actually, any one of VnA toVnF) is input and linearization as represented by the following equation(1) is performed.

θdc60=A1*Vn0+B1   (1)

FIG. 5(c) is still another example of linearization. A methodillustrated in FIG. 5(c), the neutral point potentials of a plurality ofpoints within a range of −60 [deg] to 0 [deg] are obtained. In this way,a plurality of A1 and B1 in the above equation may be prepared toimprove estimation accuracy.

FIG. 6 illustrates an example of an observation result of VnA, VnB, VnC,VnD, VnE, and VnF which are six neutral point potentials. In FIG. 6,solid lines and dashed lines are mixed, but this is for easierrecognition. If these six neutral point potentials are selected and usedat each of the rotor positions with 60 degrees, the waveform asillustrated in FIG. 7 is obtained.

FIG. 7 is a diagram in which six types of neutral point potentials areselected for each of the rotor positions with 60 degrees. In this way,by using the six types of the neutral point potentials for each of therotor positions with 60 degrees in order, a target waveform can beobserved and the rotor position can be estimated by a simple algorithm.FIG. 7 illustrates an example for a method of selecting six types of theneutral point potentials.

The above algorithm explains a basic operation of position estimationusing the neutral point potential. The basic operation uses positiondependence of the neutral point potential changed according to aposition of a magnet-magnetic flux. However, in the actual motor, thereis a case where as another factor for determining a value of the neutralpoint potential, there is current dependence which is changed by thetorque current.

FIG. 8 illustrates a result obtained by measuring the neutral pointpotential (here, VnA) while varying a value of the q axis current (thatis, a current of a torque component) to 25%, 50%, 100%, and 200%. Here,the values of the neutral point potentials when the rotor positions θdcare −60 [deg], −30 [deg], and 0 [deg] are respectively VnA0, VnA1, andVnA2. For example, VnA0_025 is the value of the neutral point potentialwhen the rotor position θdc is −60 [deg] and the q axis current is 25%.

Although degree of dependence is different depending on magnetic circuitcharacteristics of the motor, there is a case where current dependenceas illustrated in FIG. 8 may occur. Accordingly, there is a possibilitythat an accurate position is not estimated with simple fitting asillustrated in FIG. 5 and a torque specification is not satisfied; andbesides the position is not estimated.

In the present embodiment, as a factor for determining the value of theneutral point potential, by considering (1) not only position dependencechanged by the position of the magnet-magnetic flux (2) but also currentdependence changed by the torque current, the control device of thesynchronous electric motor capable of high torque driving from a zerospeed can be realized.

Next, a method of obtaining current dependence of the neutral pointpotential will be described. First, using FIG. 9(a), a relationshipbetween a permanent magnet-magnetic flux Φm and the torque current Iqduring normal driving will be described. As illustrated in FIG. 9(a), innormal driving, the torque current Iq flows in a direction orthogonal tothe magnet-magnetic flux Φm and rotation power is obtained. A magneticflux Φq due to the Iq is generated in a direction orthogonal to themagnet-magnetic flux Φm. As a result, a total magnetic flux Φ1 insidethe motor is a magnetic flux obtained by combining the Φm and the Φq.For this reason, the neutral point potential is also changed byinfluence of the Φq, but degree of influence differs depending on designof the motor.

Therefore, if measurement of the neutral point potential is performed ina state in which the Iq is energized, it is possible to obtain currentdependence, but since the motor generates rotation power by flowing theIq, the rotor must be mechanically fixed and it is difficult to easilymeasure the neutral point potential.

In the present embodiment, as illustrated in FIG. 9(b), an alternatingcurrent (referred to as AC current) flows in a direction of ±90 degreeswith respect to the magnet-magnetic flux Φm. If the Iq is not a constantcurrent of a direct current but an alternating current, an averagetorque becomes zero and a high current can flow without generatingrotation power.

Using FIG. 10, a principle for obtaining current dependence by flowingan AC current will be described. As illustrated in FIG. 10, a current ofa d axis is at zero and an AC current of a q axis flows. The current isgenerated by applying the AC voltage Vh* generated by the AC commandgenerator 20 in FIG. 1 as the q axis voltage Vq*. Since the alternatingcurrent is an AC current which does not form a rotating magnetic field,an average torque of the motor is zero.

In addition, in the present embodiment, as illustrated in FIG. 10(b), bygradually increasing an AC amplitude from zero, the rotor position ofthe motor is suppressed from moving. Although the current is thealternating current, if a high current is flowed to the q axis at once,there is a possibility that the rotor position may move due to theshock, but this is suppressed.

In addition, by gradually increasing the amplitude of the alternatingcurrent and obtaining the neutral point potential when a desired currentvalue is reached, it is possible to obtain the current value and theneutral point potential value. That is, current dependence at a certainrotor position as illustrated in FIG. 8 can be obtained.

FIG. 11 illustrates a flow of “measurement mode” for obtaining theneutral point potential also including current dependence. Processesfrom (S1) to (S14) in FIG. 11 will be described.

In (S1), the SW21 a to SW21 d are set to a “0” side and only the SW21 eis kept at a “1” side. Next, in (S2), the phase setter 17 outputsθdc=−60 [deg] and at the same time an output of the Id* generator 5 b isset to I0. The I0 is a current value necessary for moving the rotor, butas a guide, the I0 may correspond to a rated current of the PM motor 4.By setting in (S2), a DC current is generated at a position of θdc =−60[deg]. The rotor moves according to the current and stops at theposition of θdc=−60 [deg]. Next, in (S3), SW21 e is switched to “0” andthe Id* is set to zero at the same time. At this point, the DC currentis interrupted.

Next, in (S4), an AC voltage is gradually applied from an AC voltagegenerator 20 to obtain the neutral point potential VnA with respect to apeak value of the flowing AC current (waveform in FIG. 10). The VnAobtained at this time corresponds to VnA0_025 to VnA0_200 in FIG. 8since the rotor position θd is −60 [deg].

In the same manner, by setting θdc=−30[deg] by the processes from (S5)to (S8), the current dependence VnA1_025 to VnA1_200 at the rotorposition of −30 degrees is obtained. Then, by setting θdc=0 [deg] by theprocesses from (S9) to (S12), the current dependence VnA2_025 toVnA2_200 at the rotor position of 0 degree is obtained.

In (S13), an approximate function for interpolating the neutral pointpotential and the rotor position based on these neutral point potentialvalues is calculated. In the present embodiment, a linear interpolationformula as illustrated in FIG. 5 is used as the approximation functionand the coefficients A1 and B1 in the equation (1) are obtained for eachof the rotor positions and the q axis current values. As a result, it ispossible to create a table considering current dependence and to performposition estimation-operation with improved estimation accuracy.

Since the PM motor is in principle three-phase symmetric, if measurementis performed in a section of 60 degrees in an electrical angle, theresult can be applied to other phases (as illustrated in FIG. 7,waveforms for each of 60 degrees are equal to one another).

For realizing the flow of FIG. 11, it is necessary to detect the VnAwhile flowing the alternating current in (S4), (S8), and (S12). That is,when the inverter 3 generates an alternating current on the q axis, itis necessary to surely include the VA as a voltage vector. This can berealized by devising a method of generating a pulse when the pulse widthof the inverter is modulated. FIG. 12 illustrates an example of thepulse generation method.

As illustrated in FIG. 12(a), if the position of the voltage command V*is in a range of ±60 degrees, the VA is always output from the inverter.Therefore, as illustrated in FIG. 12(b), the VA is output by the PWMusing the normal triangular wave carrier. When the voltage vector VA isoutput, by measuring the neutral point potential, it is possible todetect the VnA.

However, depending on a condition, there is also a possibility that anoutput period of the VA is extremely short or the VA is not output in aregion other than ±60 degrees. In this case, “pulse shift” will beexecuted as illustrated in FIG. 12(c). The pulse shift adds correctionwithout changing a total value of original voltage commands (Vu0, Vv0,and Vw0) in a falling section of the triangular wave carrier (period Tc1in FIG. 12(c)) and a rising section (period Tc2 in FIG. 12(c)) and it ispossible to output the voltage pulse of the VA by changing the amount ofthe shift. The pulse shift is executed by the PWM generator in FIG. 1.

FIG. 13 illustrates a configuration of the position estimator 15 whichoperates during the actual operation mode. In FIG. 13, the neutral pointpotential Vn0 is input and calculation of a rotation position isexecuted by using a multiplier 152 and the adder 6 according to theequation (1). Measurement values are saved in coefficient tables 1511and 1512 so that values of A1 and B1 in equation (1) are correctedaccording to the current Iq. In the embodiment, although measurement isperformed on currents at four points (25%, 50%, 100%, and 200%), byperforming linear interpolation between the currents, it is possible todeal with a total current value. Alternatively, the number of themeasurement points may be further increased to improve accuracy of thecoefficient table.

θdc 60 is set so as to be calculated within a range of ±30 [deg],staircase waveform signals θdc 0 for each 60 degrees output by a θdreference value generator 153 are added to the θdc 60, and the estimatedphase θdc of 0 to 360 [deg] is obtained.

As described above, the control device of the synchronous electric motoraccording to the present embodiment sets the PM motor with thethree-phase stator windings Y-connected to a driving target andenergizes the AC current which does not generate rotation power by theinverter before actual driving to obtain the neutral point potential(potential of Y connection point) of the PM motor according to magnitudeof the energized current. By storing the obtained value in anon-volatile memory of the controller and executing the rotor positionestimation of the PM motor based on the value, it is possible to realizethe control device of the synchronous electric motor capable of hightorque driving from a zero speed According to the present embodiment, itis possible to easily perform automatic adjustment on parametersnecessary for the position-sensorless driving whatever PM motor is usedand realize sensorless driving of a versatile PM motor.

Second Embodiment

Next, the control device of the synchronous electric motor according tothe second embodiment of the invention will be described using FIGS. 14and 15.

In the first embodiment, a method in which it is possible to deal with aPM motor of which characteristics are unknown by the simple adjustmentalgorithm is described. In the present embodiment, the device whichsolves a problem of three-phase imbalance in each of PM motors isprovided.

In the first embodiment, for example, as illustrated in FIG. 6 and FIG.7, the adjustment algorithm is configured assuming that the neutralpoint potential of each of voltage vectors is changed equally indetection characteristics of the neutral point potential. However, inthe actual PM motor, three-phase imbalance often occurs due to amanufacturing error and material variation. In particular, the neutralpoint potential used in the present embodiment also detects an effect ofminute change in inductance of each of phases, and this method is easilyaffected by imbalance.

FIG. 14 illustrates an appearance in which measurement results of theneutral point potential for six types of voltage vectors are dispersed.These are caused by three-phase imbalance of the motor itself and alsoinclude influence due to variations of a detection circuit (virtualneutral point generator 34 in FIG. 1) of the neutral point potential.However, in the first embodiment, the variation for each of the phasescannot be compensated in the measurement mode.

In the present embodiment, in order to solve these problems, theadjustment operation is executed for each of six types of the neutralpoint potentials. In the flow of FIG. 11, the neutral point potentialfrom θd=−60° to θd=0° is measured, but the neutral point potential ismeasured in all regions of electrical angle 360°. That is, (S1) to (S13)in FIG. 11 are repeated six times to deal with 360 degrees. Using ameasurement result of the neutral point potential, coefficient tables1511B and 1512B inside a position estimator 15B illustrated in FIG. 15are prepared for each of 60 degrees and the coefficient tables areswitched using switches 154 a and 154 b.

As a result, the variation for each of three phases is compensated andit is possible to accurately operate the rotor phase θdc.

Third Embodiment

Next, the control device of the synchronous electric motor according tothe third embodiment of the invention will be described using FIG. 16.

In the first and second embodiments, as the rotor phase, the positionestimation is performed by dividing electrical angle of 360 degrees foreach of 60 degrees with reference to zero degrees. However, a waveformof the detected neutral point potential is not a target in each of60-degree periods and an error in linear approximation becomes large.Naturally, as illustrated in FIG. 5(c), although it is possible toobtain some reference points and approximate the points by a polygonalline, the process becomes complicated and operation time in themeasurement mode also becomes long.

The third embodiment according to the invention solves this problem.

FIG. 16 illustrates a comparison on principles of the embodiment (FIG.16(a)) and the embodiment (FIG. 16(b)). For example, if a 60-degreeperiod for detecting the VnA is set to a range of −60 degrees to 0degrees, a large error occurs in some parts. In the present embodiment,as illustrated in FIG. 16(b) , linearization is executed in a range of−75 degrees to −15 degrees by shifting by 15 degrees. Accordingly, thedetected VnA becomes a symmetrical waveform and the error in the linearapproximation is greatly reduced. Even if an angle of deviation is not15 degrees, the same effect can naturally be expected if the angle isnear to 15 degrees.

The flow in the measurement mode of this method only changes a settingof the phase θd in the flow of FIG. 11. For example, in the process of(S2) in FIG. 11, θdc=−60 degrees may be set as θdc=−75 degrees.Furthermore, the embodiment can be realized by deleting (S5) to (S8) andsetting θdc=0 degrees to θdc=−15 degrees in (S9). The corrected flow isillustrated in FIG. 17.

As described above, by shifting the rotor position in the measurementmode by 15 degrees, it is possible to realize sensorless driving capableof more highly accurate position estimation. Regarding three-phaseimbalance, it is possible to realize the sensorless driving in theexactly same manner by executing based on the second embodiment andshifting the moving position of the rotor in the measurement mode by 15degrees.

Fourth Embodiment

Next, the control device of the synchronous electric motor according tothe fourth embodiment of the invention will be described using FIG. 18.

As described in the previous embodiments, by obtaining the neutral pointpotential of a predetermined phase in the measurement mode, it ispossible to drive the PM motor with a high response and high quality(low torque pulsation, low loss, or the like). However, when the motoris combined with the controller, the measurement mode is executed onlyonce as an initial work and cannot deal with a temporal change of themotor characteristics. In principle, although the PM motor is a motorwith little change over time, there is a possibility that a temperatureof the motor may change from several tens degrees to approximately 100degrees during driving. Due to the temperature change, thecharacteristics of the permanent magnet attached to the rotor arechanged and the neutral point potential may fluctuate as a result. Inparticular, the measurement mode is a first-time operation mode andthere is a high possibility of adjusting under a condition in which thetemperature of the PM motor is low. On the other hand, if the PM motoris driven in the actual operation mode, the motor main body generatesheat due to a copper loss and an iron loss and there is a possibilitythat the characteristics of the motor main body may be different fromthe characteristics in the measurement mode.

Therefore, detection of the neutral point potential in the measurementmode is preferably executed as close as possible to a temperaturecondition in the actual operation mode.

Therefore, as illustrated in FIG. 18, new processes (P1) and (P2) areadded to the measurement mode algorithm (FIG. 9) of the firstembodiment. In (P1), the SW21 a to the SW21 d are temporarily set to themeasurement mode, I₀ is changed to a predetermined value and the phaseθ_(dc) is changed to 0 [deg], 120 [deg], and −120 [deg] in order in(P2), and the PM motor 4 is energized. The energization in (P2) is togenerate the copper loss due to an electric current so as to increasethe temperature of the motor to a value close to the actual operationand energization patterns maybe arbitrary. However, the energizing phasemay be changed so that the current does not concentrate in a specificphase.

If the measurement mode described in the above embodiments is executedafter energizing the PM motor 4 by (P2), the neutral point potential ata temperature condition close to the actual operation can be obtained.

As described above, according to the fourth embodiment of the invention,it is possible to obtain the neutral point potential close to the actualoperation temperature condition in the measurement mode and to improveposition estimation accuracy of the actual operation.

Fifth Embodiment

Next, the fifth embodiment of the invention will be described.

FIG. 19 is an actual diagram illustrating a driving system of thesynchronous electric motor according to the embodiment. In FIG. 19, asynchronous electric motor driving system 23 is packaged inside themotor 4 as one system. By integrating all of the synchronous electricmotor driving system 23 and the motor 4 in this manner, wiring betweenthe motor and the inverter can be eliminated. As illustrated in FIG. 19,wiring of the integrated driving system is only a power supply line tothe inverter 3 and a communication line for a rotation speed command,returning the operation state, or the like.

In the embodiment, although it is necessary to derive the neutral pointpotential of the motor 4, by integrating the motor and the drive circuitportion in this manner, the wiring of the neutral point potentialbecomes easy. In addition, since the position sensorlessness driving canbe realized, the integrated system can be extremely compact andminiaturization can be realized.

Sixth Embodiment

Next, the sixth embodiment of the invention will be described.

FIG. 20 is a hydraulic driving system used for transmission hydraulicpressure inside an automobile, brake hydraulic pressure, and the like.In FIG. 20, a component number 23 is the synchronous electric motordriving system in FIG. 19 and an oil pump 24 is attached to the motor.By the oil pump 24, hydraulic pressure of a hydraulic pressure circuitis controlled. The hydraulic pressure circuit 50 is configured toinclude a tank 51 which stores oil, a relief valve 52 which keepshydraulic pressure below a set value, a solenoid valve 53 which switchesthe hydraulic pressure circuit, and a cylinder 54 which operates as ahydraulic actuator.

The oil pump 24 generates hydraulic pressure by the synchronous electricmotor driving system 23 and drives the cylinder 54 which is thehydraulic actuator. In the hydraulic pressure circuit, by switching thecircuit by the solenoid valve 53, a load of the oil pump 24 is changedand turbulence other than the load is generated in the synchronouselectric motor driving system 23. In the hydraulic pressure circuit, aload equal to or more than several times may be applied to steady statepressure and the motor may be stopped. However, in the synchronouselectric motor driving system according to the embodiment, it ispossible to estimate the rotor position even in a stop state, so that noproblem occurs. With sensorlessness described above, since it isdifficult to apply only in a middle to high speed range, it isindispensable to relieve hydraulic pressure which becomes a heavy loadof the motor by the relief valve 52, but according to the presentembodiment, it is also possible to eliminate the relief valve 52 asillustrated in FIG. 21. That is, hydraulic pressure control becomespossible without the relief valve which is a mechanical protectiondevice for avoiding overload to the motor.

Although this embodiment is described as the hydraulic pressure controlsystem, it is also applicable to another liquid pump as well.

Seventh Embodiment

Next, the seventh embodiment of the invention will be described.

FIG. 22 illustrates a positioning device using the motor and an entireblock configuration of the positioning device. In FIG. 22, thepositioning device 70 is connected as a load of the motor 4. Here, anIq* generator 1E functions as a speed controller. In addition, a speedcommand ωr* is provided as an output of a position controller 71 whichis an upper control block. A subtractor 6E compares the speed commandωr* with an actual speed ωr and operates the Iq* so that deviationbecomes zero. The positioning device 70 is a device using, for example,a ball screw and is adjusted by the position controller 71 so that theposition is controlled to be a predetermined position θ*. The positionsensor is not attached to the positioning device 70 and the positionestimation value θdc in the controller 2 is used as it is. Thereby,there is no need to attach the position sensor to the positioning deviceand position control can be performed.

Although the invention has been specifically described based on theembodiment, the invention is not limited to the above embodiment, andvarious modifications can be made without departing from a gist thereof.

As described above, the invention is a technology for constructing thecontrol device of the synchronous electric motor on the premise of theposition-sensorlessness and a driving system using the control device.An application range of this motor can be used as a conveyor, anelevator, an extruder, a machine tool as well as rotational speedcontrol of a fan, a pump (hydraulic pumps, water pumps), a compressor, aspindle motor, an air conditioner, and the like.

The disclosure content of the following priority application isincorporated herein as a quotation. Japanese Patent Application No.2015-199040 (Oct. 7, 2015)

REFERENCE SIGNS LIST

1 . . . Iq* generator

2 . . . controller

3 . . . inverter

31 . . . DC power supply

32 . . . inverter main circuit

33 . . . gate driver

34 . . . virtual neutral point potential generator

35 . . . current detector

4 . . . PM motor

5 . . . Id* generator

6 . . . adder

6, 7 . . . d axis current controller IdACR

8 . . . q axis current controller IqACR

9 . . . dq inverse converter

10 . . . pulse width modulation unit

11 . . . current reproducer

12 . . . dq converter

13 . . . neutral point potential amplifier

14 . . . sample/holder

15 . . . position estimator

16 . . . speed operator

17 . . . phase setter

18 . . . estimation parameter setter

19 . . . zero generator

20 . . . AC command generator

21 . . . switch

1. A control device of a synchronous electric motor, which controls thesynchronous electric motor using an inverter, the device comprising: thesynchronous electric motor with three-phase stator windings Y-connected;a detection unit that detects a neutral point potential which is apotential at a Y connection point; and the inverter that drives thesynchronous electric motor, wherein the control device of thesynchronous electric motor includes a measurement mode in which theneutral point potential is detected in a state in which the synchronouselectric motor is energized by an AC current and controls thesynchronous electric motor based on a value of the neutral pointpotential detected in the measurement mode.
 2. The control device of thesynchronous electric motor according to claim 1, wherein after thesynchronous electric motor is energized by a direct current to move arotor, the neutral point potential is detected by the AC current in themeasurement mode.
 3. The control device of the synchronous electricmotor according to claim 2, wherein energization by the direct currentand detection of the neutral point potential by the AC current arerepetitively executed at a plurality of positions of the rotor.
 4. Thecontrol device of the synchronous electric motor according to claim 3,wherein a plurality of movement ranges of the rotor by the energizationby the direct current are a plurality of points within a range of atleast 60 degrees with respect to an electrical angle of the synchronouselectric motor.
 5. The control device of the synchronous electric motoraccording to claim 3, wherein a plurality of movement ranges of therotor by the energization by the direct current are a plurality ofpoints within a range of 360 degrees with respect to the electricalangle of the synchronous electric motor.
 6. The control device of thesynchronous electric motor according to claim 3, wherein a directcurrent energization phase to the synchronous electric motor includes atleast one phase of a plurality of points at intervals of 60 degrees withreference to a phase shifted by 15 degrees with respect to a zerodegree, where a position of a U-phase stator winding is defined as thezero degree of an electrical angle phase.
 7. The control device of thesynchronous electric motor according to claim 1, wherein the AC currentfor energizing the synchronous electric motor energizes the synchronouselectric motor while an amplitude of the AC current being changed. 8.The control device of the synchronous electric motor according to claim7, wherein a maximum amplitude value of the AC current for energizingthe synchronous electric motor is at least equal to or larger thanmagnitude of the current during normal driving of the synchronouselectric motor.
 9. The control device of the synchronous electric motoraccording to claim 7, wherein the amplitude of the AC current forenergizing the synchronous electric motor gradually increases from zero.10. The control device of the synchronous electric motor according toclaim 1, wherein when the measurement mode is executed, after thesynchronous electric motor is energized in advance, the neutral pointpotential is detected.
 11. The control device of the synchronouselectric motor according to claim 1, wherein during driving when thesynchronous electric motor is normally driven after the neutral pointpotential is detected in the measurement mode, the neutral pointpotential is detected to detect the neutral point potential by applyinga voltage pulse for neutral point potential detection by the inverter, arotation position of the synchronous electric motor isestimation-operated from a value of the detection and a value of thedetected neutral point potential in advance, and the synchronouselectric motor is driven based on the rotation position.
 12. The controldevice of the synchronous electric motor according to claim 11, whereinthe voltage pulse for the neutral point potential detection outputduring normal driving of the synchronous electric motor is applied to atleast one of an upward period and a downward period of a triangular wavecarrier wave with respect to the triangular wave carrier wave when apulse width of the inverter is modulated to detect the neutral pointpotential, and the rotation position is estimated based on a value ofthe detection.
 13. The control device of the synchronous electric motoraccording to claim 11, wherein rotor position estimation of thesynchronous electric motor is operated by a function based on thedetected neutral point potential in advance.
 14. An integrated electricmotor system comprising: the control device of the synchronous electricmotor according to claim 1; and a rotor and a stator of the synchronouselectric motor, driven by the control device of the synchronous electricmotor, stored in a common housing.
 15. A pump system comprising: tocontrol device of the synchronous electric motor according to claim 1;the synchronous electric motor driven by the control device of thesynchronous electric motor; and a pump for liquid driven by thesynchronous electric motor.
 16. A positioning system which moves anobject with the control device of the synchronous electric motoraccording to claim 1; and the synchronous electric motor driven by thecontrol device of the synchronous electric motor and the electric motorand controls a position of the object.